My old vertical tuner can’t withstand QRO power levels as the tuner is limited to approx 150W. I am therefore working on a new multiband vertical for QRO operation. Its best to try to be finished before winter sets in(soon approaching as I write this blog post). Instead of using traps, I will tune the antenna like a Marconi type antenna over a ground plane with switchable or tunable L/C networks down at the feed point. The challenge with multiband antennas that is going to cover all the 40, 30, 30, 20, 17, 15, 12 and 10m bands, is that there will be frequency ranges where the real impedance is very high. A high real impedance is not possible to tune out with L or C and also difficult to match with a L network. It can be fed and matched with a tapped parallel network, but there will be high voltages present and vacuum capacitors will be needed for QRO operation. L networks are easier on most bands and I try to use only one HV capacitor in one parallel network for 18 MHz (at least that is the plan). The trick is to have a proper length radiator that is tuned so that the impedance peaks will lay outside the ham bands of use. At least one band will have high impedance, but it should be possible to have fairly low impedance above that band. I did a 4NEC numeric antenna simulation to investigate the expected impedance range before sizing the radiator in the real life. What to look for is the zero phase transitions (look at the pink curve above). The first zero phase transition is the quarter wave resonant point simulated in 4NEC to be around 8,5 MHz. The next zero phase transition is around 17 MHz. This is the half wave resonant point. The impedance is very high at this frequency. Then there is a zero phase transition around 25,5 MHz. This is the 3/2 lambda resonant point. Here the impedance is low again. From the simulation graph it can be seen that 7, 10, 14, 21, 24 and 28 MHz will be possible to match with a L network. 18 MHz will have to be voltage fed because the impedance is very high.
To verify the simulations made with the Numerical Electromagnetic Code (4NEC) simulator I did some vector network analyzer measurements in the feed point end of the self supporting fiberglass mast that supports the vertical. The VNA S11 plot can be seen on the PC. The VNA unit is placed inside the tuner enclosure. (The ground plane is buried and is relatively extensive). The impedance peak of the half wave resonant point can be seen on the PC. However, there were some unexpected effects that affected the VNA measurements. I suspect that the master calibration was not good. Will have to look at that later. (The blue plastic sheet placed on the ground is laid there to be able to more comfortably work on the ground without becoming wet and dirty. The gray “ring” to the right is a concrete support for my soldering iron (ELRA ca. 1980 model still in good shape). I use a chair as a “PC support” to avoid placing the laptop on the ground. Cables to the house and control cables are routed below the surface in tubing.
I Recently bought some DX-Wire DXW-174 cable from http://www.dx-wire.de/brit/ I was eager to check out if the cable meets the published specifications. (See below picture in yellow. This is the published specifications from the manufacturers DX-WIRE’s website )
To verify the published specs, I took a roll of 100m and did a S21 measurement with my vector network analyzer. The VNA was calibrated with open, short, 50+jØ, crosstalk, thru, completely open loads. The S21 plot with dB scale and 5dB/division is shown below.
Results: the loss at 100kHz measures 1,29dB and the loss at 28 MHz measures 13,51dB.
Conclusion: this is almost exactly the spec that DX-Wire states on his webpage. I am satisfied with the accurate data published by this supplier. I therefore can recommend his product. However be aware of that 100m on HF with this cable will give you far too high loss even at 80meter. 10-15 meters should be no problem though. The loss on lower frequencies is lower than other cables made for higher frequencies with no solid copper core.
Experiment 1: I did an S11 analysis with my VNA on a 500pF transmitting capacitor in parallel with an airwound copper coil of good cross section. The question is how to interpret the first picture. The voltage over the parallel circuit is in phase with the drive voltage and almost as high in amplitude as the drive voltage when there is a real component present. (Blue trace, left image). The drive voltage cannot drive a current when the circuit is in steady state oscillation since the circuit has almost the same voltage over it as the drive voltage itself. Low voltage differential, results in low current. This is the definition of high impedance. Low in frequency the resultant voltage is not in phase however the voltage amplitude close to the resonance is almost as high as the drive voltage. The circuit presents a inductive load. Above the frequency there is a change from inductive to capacitive load. (Red trace). It looks like the phase changes abruptly, but I interpret it as a high impedance that occurs only due to a change from high inductive to high capacitive voltage. The phase goes “over the top” but only since someone defined capacitive reactance as minus. The impedance is still high and almost real close to resonance. |Z| represents the resistance to AC current in this context. It is the length of the R+jX vector.
Experiment 2: after that I did a S21 sweep of the same components in a series notch configuration as per VE2AZX’s paper http://www.arrl.org/files/file/QEX_Next_Issue/Jan-Feb_2012/QEX_1_12_Audet.pdf to try to determine the Q (of the coil primarily – I assumed the Q of the capacitor very high compared to the coils Q).
The second picture gives the notch depth so the Q can be calculated as per VE2AZX’s QEX paper http://www.arrl.org/files/file/QEX_Next_Issue/Jan-Feb_2012/QEX_1_12_Audet.pdf
Note1: the picture to the right is of the S21 test, not of the parallel circuit and the frequency may have changed slightly as the coil were changed between the two experiments if I recall correctly.
Note2: don’t worry about the long leads. This is an exercise in understanding the design and operation of antenna trap circuits and loss in reap circuits due to resistive losses in the inductor primarily. The leads would be almost as long in a real trap.
I wanted to check the load that a ferrite core with a secondary winding presents to the common mode current carrying conductor (the outside of a coax) when set up as a current transformer. The coax runs thru center, the secondary winding could be one or several turns loaded by a R+jø load. (+jø load but the windings of the secondary will give some +jx component).
I did the following three S11 measurements with the VNA (M1, M2, M3):
M1) No secondary winding is present but the single turn is running thru the core
M2) A shorted two turn secondary and an open two turn secondary (winding is pulsed in and out off to be able to better detect a difference). The time domain is captured by the slow scan and sampling rate of the VNA.
M3) Only the primary winding is attached to the calibrated S11 measuring plane. (to measure the self inductance of the single turn in itself. This is an air-core measurements.)
Above is a similar test setup. The core is an “unknown” Amidon toroid core. The red conductor is the simulated coax common mode current path (a one turn loop). The green conductor is the secondary winding. The load could made by paralleling several resistors in series with the secondary (the blob on the left side, right picture). Note that the measurements below were done with a one turn loop and a short – no resistor. The measurement device is a VNA from DG8SAQ calibrated by O S L references in the S11 measuring plane (the SMA in the end of the coax from the TX port).
The left picture shows measurement 1) The mid picture shows measurement 2) the right picture shows measurement 3)
Edited: What is interesting to see is that the image to the left shows that this core has some resistive loss as can be seen on the blue trace. The Q is quite low. When there is a secondary winding present, the loss is shorted out but the inductancechanges since the inductance of the secondary is reflected into the S11 measurement plane. On the right image it can be seen that there is no resistive loss and a linear inductive reactance caused by the air core inductor (no appreciable drop off or frequency dependent effects, in the measuring frequency range)
What this tells me is that this core setup probably is not too well suited as a high current measuring setup for frequencies above 160m because it will affect the measuring circuit too much. Not in terms of the R element but because of the +jX element. The R is low to the RF current passing thru the core (in the common mode), but the +jX element will present a reactance to the RF current and thereby giving you lower current than should be expected without the core. My analysis says that another core type should be selected or that a frequency compensation technique should be used. Alternatively that a lower turns ratio should be tried. However looking at the plot to the left, it can be seen that below 30 Mc*s^-1 the +jX component is too high. Perhaps this core has a too large permeability and that a lower permeability core should be used. The primary turn with secondary loading should give a low reactance on the primary. The voltage given over Rt would then be lower but the gain of the detector could be adjusted. (I may be wrong). Please comment if you have comments or suggestions.
Today I did some further measurements on the same Fair-Rite material as the other post (Fair-Rite 2643167851). I did the measurements with a calibrated open, short, 50 ohm load S11 measurement. The green trace is the resistive part of the impedance, the red trace is the reactive part of the impedance and the blue trace is the unloaded Q (ratio stored energy in the magnetic field in the core and windings to lost energy in the effective series resitance). As expected, the Q is one when the green and red traces cross each other. This material is not useable as a regular coil for energy storage (high Q, filters and such) over approx 15 Mhz. Below 15 MHz the Q can be quite high, however. This core is probably very good as a dampening material for RFI applications as GM3SEK has indicated. For High power operation when coax common mode current causes problems I think this core may be good. The reason is that the resistive component (real component) of the impedance is dominant. This means that even if a capacitive reactance cancels the inductive reactance, the resistive part of the impedance is still always present. This can be seen from the green trace above. Center of the plot is approx 300Mc/sek. It also rises with QRG up around 450 Mc/sec. At 2 meters and 70 centimeters wavelength it looks like even one turn on a coax (in a low impedance point of the coax!) this choke will do some good. At HF, with more turns it is possible to achieve 1000 ohms over a fairly large range.
After reading the nice publication by GM3SEK (http://www.ifwtech.co.uk/g3sek/in-prac/inpr1005_ext_v2.pdf) about the Fair Rite ferrite matrial that gives a high resistive component when used for RF choking applications I wanted to do some measurements myself vith my VNA on that material used as a RF choke.
Above is a three choke setup
The blue trace above is the real Z ( R ) plotted from 0-30 MHz. The scale is 480 ohms/div. As you can see the material gives a resistance over a fairly large BW well over 1000 ohms peaking around 5K ohms resistive at the low frequency range. However over approx 18 Mhz, the resistive component is not that great. For QRO operation on 15 meter and 10 meter some further measures should be taken if RF currents are high.
Here is a three stage RF-choke setup. First the three element RF-choke, then a one element RF-choke and a two element RF-choke with the material stacked on top.
Note that the scale here is 620 ohms/div. The measurements show that the three in line setup didnt change the resistive part of the choke a lot. This may indicate that for QRGs over 18-20 MHz, a material with more loss in that frequency range could be found. I am later going to experiment with different winding diameters and cable diameters to see if that will change the resistive elements in the upper parts of the HF spectrum.
I wanted to test a simple airband antenna design since I had a roll of balanced feedline laying around in my shack doing nothing useful. First I had to measure the velocity constant of the feedline with my MFJ-259 to be able to come to an estimate of the required length of the matching section. I could also have used my vector network analyzer (VNA) to do that by the way. To measure without interference from coupling to adjacent objects I did the measurement with the cable hanging out from my balcony as you can see in the left picture. The MFJ-259 was connected in the end and held by hand (that is a benefit of the battery operated MFJ 259 even if the instrument is not of the most accurate on the market). I wanted to make two antenna segments folded over each other. Therefore the top of the feedline is shorted and the currents will be in phase if the antenna is of a proper length. The matching section is a shorted line section that is tapped by the transmission line. The coil on the coax is a choke (I haven’t done any measurements on that choke yet by the way).
The procedure I used to find the velocity factor of the balanced transmission line was to first measure a length of the feedline with a tape measure. Then I connected the MFJ-259 and found the frequency where the lowest reactance could be measured. (See pic two from left on the upper row above. You can see that the X is very low). This was done in the mode of the MFJ-259 where it is possible to measure both R and X. This is the quarter wave frequency of the line when the wave propagates in the line – not in the free air (“ether”). Then I calculated that frequency back to the wavelength with y=300/f. I then divided the tape measure length by the calculated length and came to a velocity factor of 0,89. This is the ratio of the wavelength in free air and the wavelength in the transmission line. This is directly related to the propagation speed of the line when it operates in transmission line mode. From that I calculated the required length of the matching transformer and the approximate tapping point on that transformer to reach 50 ohms. Please note that you cannot use the velocity factor of the transmission line to calculate the required length of the antenna (only the matching section), since the RF currents on the two folded legs on the antenna are in phase and therefore the one lead is coupled to the ether and not to the other lead.
The picture above shows the SWR as measured with my vector network analyzer from DG8SAQ. The markers on the right side shows a 1:2 SWR bandwidth of 118,5 to 128,6 Mc/s which is OK. The reference level is 1:1 SWR. This level is lifted one division for clarity. (I think the Mc/s is a cool way to express frequency by the way.)
Here my DG8SAQ1 kc/s to 1,3 Gc/s VNA is shown. It is connected to my PC via USB.
Conclusion: a combination of the MFJ-259, the DG8SAQ vector network analyzer, some balanced line and some coax can be used to make a good collinear airband antenna in less than one our at a cost of a few dollars. The antenna was screwed to a wooden section of my roof by a small screw by the way and can be removed in approx 2 minutes.